Report

Solving Op Amp Stability Issues Part 2 (For Voltage Feedback Op Amps) Tim Green & Collin Wells Precision Analog Linear Applications 1 Stability Tricks and Rules-of-Thumb Loop Gain Bandwidth Rule: 45 degrees for f < fcl Aolβ (Loop Gain) Phase Plot 180 fcl 90o Phase Shift at fcl fp1 (degrees) 135 Frequency (Hz) 90 10 45o “Phase Buffer” away from 180o Phase Shift 45 0 -45 100 1k fp2 10k fz1 100k 1M 10M 90o Phase Margin at fcl Loop Stability Criteria: < -180 degree phase shift at fcl Design for: < -135 degree phase shift at all frequencies < fcl Why?: Because Aol is not always “Typical” Power-up, Power-down, Power-transient Undefined “Typical” Aol Allows for phase shift due to real world Layout & Component Parasitics Prevent excessive ringing due to phase margin dip near fcl 3 Frequency Decade Rules for Loop Gain For 45O Phase Buffer away from 180O Phase Shift fp1 fp1 fp2 fz1 fp3 Aol pole pole --------- 1/b --------zero pole Loop Gain pole pole pole zero 100 Aol A (dB) 80 Loop Gain View: Poles: fp1, fp2, fz1; Zero: fp3 Rules of Thumb for Good Loop Stability: Place fp3 within a decade of fz1 fp1 and fz1 = -135° phase shift at fz1 fp3 < decade will keep phase from dipping further Place fp3 at least a decade below fcl Allows Aol curve to shift to the left by one decade 60 fcl 40 Rn fp3 1/Beta 20 Cn RF fp2 fz1 + 0 VIN 1 10 100 1k 10k 100k 1M 10M VOUT RI + CL - Frequency (Hz) VOUT/VIN Note locations of poles in zeroes in Aol and 1/β plots 4 Frequency Decade Rules for Loop Gain Phase Plot Prediction Phase Shift (deg) Phase Shift (deg) Individual Pole & Zero Plot +90 fp3 +45 0 fp1 -45 -90 fz1 1 10 fp1 fp2 fz1 fp3 +180 Aol pole pole --------- 100 1/b --------zero pole Loop Gain pole pole pole zero fp2 1K 10K Frequency (Hz) 100K 1M fcl At fcl: Phase Shift = 135O Phase Margin = 45O 100K 1M Final Plot 10M +135 +90 +45 0 1 10 100 1K 10K 45O Frequency (Hz) “Phase Buffer” Note locations of poles in zeroes in Aol and 1/β plots 10M 5 Op Amp Circuits & Second Order Systems 100 fp1 80 RI RF 4.7k 4.7k Aol A (dB) VOUT 60 + + VIN 40 - 1/Beta fcl 20 fp2 0 1 10 100 1k 10k 100k 1M 10M Frequency (Hz) Most Op Amp Circuits are adequately analyzed, simulated, and real world tested using well-known second order system response behavior. Most Op Amps are dominated by Two Poles: Aol curve shows a low frequency pole, fp1 Aol curve also has a high frequency pole, fp2 Often fp2 is at fcl for unity gain This yields 45 degrees phase margin at unity gain 6 Control Loop - Second Order System R(s) G(s) C(s) 2 C(s) ωn 2 R(s) s 2ωn s ωn 2 w here: ωn naturalfre que ncy(rad/s ) ζ dam pingratio Clos e dLoopRe s pons e(Ste p and AC) : Unde rdam ped : 0 ζ 1 M arginallyStable : ζ 0 CriticallyDam pe d: ζ 1 Ove rdam pe d: ζ 1 7 Closed Loop Peaking in AC Frequency Sweep vs Phase Margin Phase Margin AC Peaking @ wn * 90° -7dB 80° -5dB 70° -4dB 60° -1dB 50° +1dB 40° +3dB 30° +6dB 20° +9dB 13° +10dB 10° +14dB * Phase Margin: 1) Measure closed loop AC peaking in dB 2) Find dB peaking on graph above and read closest damping ratio, 3) Use damping ratio, , to find phase margin using Slide 75 graph 8 From: Dorf, Richard C. Modern Control Systems. Addison-Wesley Publishing Company. Reading, Massachusetts. Third Edition, 1981. Transient Real World Stability Test RI RF Volts +VS VOUT VOUT VOffset -VS VIN 1kHz 50mVPP + RL IOUT + time - Test Tips: Choose test frequency << fcl “Small Signal” AC Output Square Wave (1kHz usually works well) Adjust VIN amplitude to yield output <50mVpp Worst case is usually when VOffset = 0 Largest Op Amp RO (IOUT = 0) Use VOffset as desired to check all output operating points for stability Set scope = AC Couple & expand vertical scope scale to look for amount of overshoot, undershoot, ringing on VOUT small signal square wave Use X1 Scope Probe on VOUT for best resolution 9 2nd Order Transient Curves * 25% * Overshoot (%) 10 15 20 25 30 35 40 45 50 Phase Margin (deg) 57.5 50 42.5 40 35 32.5 30 27.5 25 * Phase Margin: 1) Measure closed loop transient overshoot (%) 2) Find overshoot on graph above and read closest damping ratio, 3) Use damping ratio, , or Overshoot (%) to find phase margin using Slide 75 graph 10 From: Dorf, Richard C. Modern Control Systems. Addison-Wesley Publishing Company. Reading, Massachusetts. Third Edition, 1981. 2nd Order Damping Ratio, Overshoot, Phase Margin 1. 2. 3. 4. Overshoot = 40%, Phase Margin = 30 degrees Start at % Overshoot on x-axis Read up to “Damping Ratio” on “Percent Maximum Overshoot” curve Read Across to Damping Ratio on y-axis Use Damping Ratio to read across to “Phase Margin” curve. Overshoot (%) 10 15 20 25 30 35 40 45 50 = 0.3 = 0.3 fM = 30deg Phase Margin (deg) 57.5 50 42.5 40 35 32.5 30 27.5 25 40% = Overshoot 11 When RO is really ZO!! OPA627 has RO OPA2376 has ZO Capacitive Inductive Resistive OPA627 OPA2376 Resistive Note: Some op amps have ZO characteristics other than pure resistance – consult data sheet / manufacturer / SPICE Model 12 Open Loop Output Impedance – SPICE Measurement SPICE Zo Te s t: L1 1TH Run SPICE AC Analys is IG1 is AC Curre ntGe ne rator IG1 DC Value 0A for unloade d Zo Zo Vout Conve rt Vo ut (dB) to Vout (Logarithmic) Zout (ohm s ) Vout (Logarithmic) Vee 15V C1 1TF - + Vout + Vcc 15V T IG1 U1 OPA627E 55.5 Ro (ohms) 55.0 54.5 54.0 53.5 1 10 100 1k 10k 100k Frequency (Hz) 1M 10M 100M 13 Summary for Stability For Stability Loop Gain Analysis all we need is: 1) 2) 3) 4) Aol – from op amp data sheet or macromodel 1/b – basic by application, modified for stability Z_Load – given by application Zo – Op Amp open loop, small signal AC output impedance from op amp data sheet or macromodel Stability General Comments: 1) 2) 3) 4) 5) 6) 7) 8) Stability by modifying 1/b will decrease closed loop bandwidth Stability compensation can slow large signal response (charging of caps) – check it Simulate AC Transfer function (Closed Loop AC Response) as final check Simulate Small Signal Transient Response as final check DC operation in the lab does not guarantee stability Marginal stability can cause undesired overshoot and ringing DC circuits can get real world transient inputs from supplies, inputs, or output That ringing in your circuit is not your Grandmother’s dial telephone T 64.94m VOUT 33.64m 100u 150u 200u Time (s) 250u 300u 14 Acknowledgements A special thanks to Jerald Graeme, whom we honorably dub “The Godfather of 1/b” for his work at Burr-Brown Corporation in research and writing about Op Amp Stability using 1/b. Jerald Graeme Brief Biography: From: http://electronicdesign.com/analog/jerald-graeme When ICs and op amps were separate devices, Jerald Graeme was among the first to develop a combined IC op amp while at Burr-Brown, in a 1968 team effort with Motorola. He designed many more op amps and video amplifiers whose precision, high speed, or low drift amplification made them a very useful component in a variety of analog applications. Nine U.S. patents and numerous foreign counterparts resulted from these designs. The internationally acknowledged authority on electronic amplifiers wrote five very popular books about op amps, the latest being Photodiode Amplifiers: Op Amp Solutions and Optimizing Op Amp Performance. The latter, subtitled "A new approach for maximizing op amp behavior in circuit designs without extensive mathematical analysis," offers design equations and models that reflect real-world op amp behavior and makes analysis of difficult-looking configurations easy. Graeme's earlier books are: Op Amps: Design and Application, Designing with Operational Amplifiers, and Amplifier Applications of Op Amps. He expects signal processing with op amps to be the domain of digital devices, but they will still require an analog interface to integrate with real-world items like process control or avionics. Jerald Graeme Books: http://www.amazon.com/Jerald-G.-Graeme/e/B001HO9X60 15 Acknowledgements Op Amp Stability Series Tim Green, Senior Analog Applications Engineer Texas Instruments http://www.engenius.net/site/zones/acquisitionZONE/technical_notes/acqt_050712 16 Appendix 17 Appendix Index 1) Op Amp Output Impedance 2) Pole and Zero: Magnitude and Phase on Bode Plots 3) Dual Feedback Paths and 1/b 4) Non-Loop Stability Problems 5) Stability: Riso (Output Cload) 6) Stability: High Gain and CF (Output Cload) 7) Stability: CF Non-Inverting (Input Cload) 8) Stability: CF Inverting (Input Cload) 9) Stability: Noise Gain Inverting & Non-Inverting (Output Cload) 10) Stability: Noise Gain and CF (Output Cload) 11) Stability: Output Pin Compensation (Output Cload) 12) Stability: Riso w/Dual Feedback (Output Cload) – Zo, 1/b, Aol Technique 13) Stability: Discrete Difference Amplifier (Output Cload) 18 Appendix Index Appendix No. Title 1 Op Amp Output Impedance Pole and Zero: 2 Magnitude and Phase on Bode Plots 3 Dual Feedback Paths and 1/β 4 Non-Loop Stability Problems Description/Stability Technique Zo vs Zout difference and datasheet curves Closed loop magnitude and phase shifts of a signal frequency due to poles and zeroes on a Bode Plot How to avoid problems when using dual feedback paths for stability compensation Oscillations and causes not seen in loop gain stability simulations When to use the Stability Technique Zo is a key parameter for stability analysis Magnitude and phase shift at a frequency of interest for closed loop poles and zeroes Key tool in analyzing op amp circuits that use dual feedback for stability Check all designs to avoid oscillations that do not show up in SPICE simulation Output Cload, Note: accuracy of output is dependent upon load current 5 Riso (Output Cload) Stability: Isolation resistor with feedback at op amp 6 High Gain and CF (Output Cload) Stability : High gain circuits and a feedback capacitor 7 CF Non-Inverting (Input Cload) Stability : Non-inverting gain and feedback capacitor Output Cload, closed loop gain >20dB Input Cload, non-inverting gain, large value input resistor 8 Stability: Inverting gain and feedback capacitor Input Cload, non-inverting gain, large value input resistor, photodiode type circuits 9 CF Inverting (Input Cload) Noise Gain Inverting and Non-Inverting (Output Cload) 10 Noise Gain and CF (Output Cload) Stability: Noise Gain added by input R-C network Stability: Noise Gain (input R-C) and feedback capacitor 11 12 Output Pin Compensation (Output Cload) Riso w/Dual Feedback (Output Cload) - Zo, 1/β, Aol Technique Stability: Series R-C on op amp output to ground Stability: Isolation resistor with two feedback paths analysis by Zo, 1/β, and Aol technique 13 Discrete Difference Amplifier (Output Cload) Stability: Balanced use of noise gain (series R-C) Output Cload, closed loop gain <20dB Output Cload, Loaded Aol has a second pole located >20dB Output Cload, no access to -input, monolithic, integrated difference amplifiers, complex feedback where not practical to use -input Output Cload, some additional Vdrop across isolation resistor is okay, accurate Vout at load Output Cload, difference amp configuration, any closed loop gain 19 1) Op Amp Output Impedance Open Loop (ZO) & Closed Loop (ZOUT) Op Amps and “Output Resistance” Definition of Terms: RO = Op Amp Open Loop Output Resistance ROUT = Op Amp Closed Loop Output Resistance RF RI RO -IN RDIFF VFB xAol + VE VOUT VO IOUT - 1A + +IN Op Amp Model ROUT = RO / (1+Aolβ) ROUT = VOUT/IOUT 21 From: Frederiksen, Thomas M. Intuitive Operational Amplifiers. McGraw-Hill Book Company. New York. Revised Edition. 1988. Derivation of ROUT (Closed Loop Output Resistance) 1) b = VFB / VOUT = [VOUT (RI / {RF + RI})]/VOUT = RI / (RF + RI) 2) ROUT = VOUT / IOUT 3) VO = -VE Aol 4) VE = VOUT [RI / (RF + RI)] 5) VOUT = VO + IOUTRO 6) VOUT = -VEAol + IOUTRO Substitute 3) into 5) for VO 7) VOUT = -VOUT [RI/(RF + RI)] Aol+ IOUTRO Substitute 4) into 6) for VE 8) VOUT + VOUT [RI/(RF + RI)] Aol = IOUTRO Rearrange 7) to get VOUT terms on left 9) VOUT = IOUTRO / {1+[RIAol/(RF+RI)]} Divide in 8) to get VOUT on left 10) ROUT = VOUT/IOUT =[ IOUTRO / {1+[RIAol / (RF+RI)]} ] / IOUT Divide both sides of 9) by IOUT to get ROUT [from 2)] on left 11) ROUT = RO / (1+Aolβ) Substitute 1) into 10) ROUT = RO / (1+Aolβ) 22 ROUT vs RO RO does NOT change when Closed Loop feedback is used ROUT is the effect of RO, Aol, and β controlling VO Closed Loop feedback (β) forces VO to increase or decrease as needed to accommodate VO loading Closed Loop (β) increase or decrease in VO appears at VOUT as a reduction in RO ROUT increases as Loop Gain (Aolβ) decreases 23 When RO is really ZO!! OPA627 has RO OPA2376 has ZO Capacitive Inductive Resistive OPA627 OPA2376 Resistive Note: Some op amps have ZO characteristics other than pure resistance – consult data sheet / manufacturer 24 With Complex ZO, Accurate Models are Key! L1 1TH SPICE Te s tof Op Am pM acro Zo : V2 2.5V C1 1TF 4 3 2 - Vout + + 1 5 U1 OPA2376 V1 2.5V T IG1 0 Run SPICE AC Analys is IG1 is AC Curre ntGe ne rator IG1 DC Value 0A for unloade d Zo Zo Vout Conve rt Vout (dB) to Vout (Logarithmic) Zout (ohm s ) Vout (Logarithmic) 1.00k Impedance (ohms) OPA2376 100.00 10.00 400uA Load 1.00 100.00m 10 100 1k 10k 100k Frequency (Hz) 1M 10M 25 Some Data Sheets Specify ZOUT NOT ZO Recognize ROUT instead of RO: ROUT inversely proportional to Aol ROUT typically <100 at high frequency ROUT is Inve rs eof Aol : RO 1 Aolb 1 Aol ROUT ROUT This is really ZOUT or ROUT! TLC082 1 TLC082 3 2 2 3 1 26 Some Data Sheets Specify ZOUT NOT ZO Point frequency Aol ---------(Hz) (dB) 1 10M 0 2 100k 40 3 10k 60 ROUT (AV=1) Datasheet (ohms) 100 2 0.2 ROUT (AV=1) Computed (ohms) RO = 200 2 0.2 TLC082 Com puteR O fromR OUT w hereAolb 1 (0dB) : 1 2 R OUT 100Ω for A V 1, f 10MHz, Aolb 1 (0dB) RO 1 Aolβ R 100 O RO 200 1 1 ROUT 3 27 2) Pole and Zero: Magnitude & Phase on Bode Plots 28 Formulae for Pole and Zero Calculations Ze roM agnitudefor f w he re: Pole M agnitudefor f w he re: f fre que ncyof s ignal f fre que ncyof s ignal fz fre que ncyof pole fp fre que ncyof pole Magnitude( dB) 20 LOG10 Pole Phase Shift for f w here: f frequencyof signal fp frequencyof pole Phase( deg ) tan1( f ) fp 1 f2 1 2 fp Magnitude( dB) 20 LOG10 1 f2 fz2 Ze roPhas e Shift for f w he re: f fre que ncyof s ignal fz fre que ncyof pole f Phase( deg ) tan1( ) fz 29 Closed Loop Gain: Magnitude and Phase Clos e dLoopM agnitudeand Phas e for fre que ncyf : w he re: f fre que ncyof s ignal fp1 fre que ncyof pole 1 fp2 fre que ncyof pole 2 fz1 fre que ncyof ze ro1 fz2 fre que ncyof ze ro2 K DC Gain in V/V Magnitude( dB) 20 LOG10 K 20 LOG10 Phase(deg) tan1( 1 1 f2 f2 20 LOG10 20 LOG10 1 2 20 LOG10 1 f2 f2 f z1 f z22 1 1 f p12 f p22 f f f f ) tan1( ) tan1( ) tan1( ) f p1 f p2 f z1 f z1 30 Closed Loop Gain: Magnitude and Phase 1 1 10.00974485Hz 2 R1 C1 2 1k 15.9F 1 1 fp2 1.004766055kHz 2 RF CF 299k 1.6nF fp1 CF 1.6nF RF 99kOhm V2 2.5V RI 1kOhm U1 OPA364 Clos e dLoop M agnitudeand Phas e for fre que ncyf : + + VOUT + w he re: f fre que ncyof s ignal R1 1kOhm VIN V1 2.5V fp1 fre que ncyof pole 1 fp2 fre que ncyof pole 2 C1 15.9uF K DC Gain in V/V Magnitude( dB) 20 LOG10 K 20 LOG10 Phase(deg) tan1( 1 1 20 LOG10 2 f f2 1 1 fp12 fp22 f f ) tan1( ) fp1 fp2 Magnitude( dB) 20 LOG10 (100) 20 LOG10 Phase(deg) tan1( 1 1 20 LOG10 2 f f2 1 1 (10.00974485)2 (1.004766055e3)2 f f ) tan1( ) 10.00974485 1.004766055e3 31 Closed Loop Gain: Magnitude and Phase T 40 20 [B] Gain : 0 VOUT B:(3.924588k; -24.084018) [A] Gain (dB) -20 Phase : VOUT B:(3.924588k; -163.475026) Gain : -40 VOUT A:(92.20173; 20.624635) Phase : -60 VOUT A:(92.20173; -89.067371) -80 -100 -120 -140 0.00 Phase [deg] -45.00 -90.00 -135.00 -180.00 1 10 100 1k 10k Frequency (Hz) 100k 1M 10M 32 Spice Compared with Calculated Analysis SPICE AC Analysis: For best accuracy use highest resolution i.e. maximum “Number of Points” f (Hz) ---------9.22017300E+01 3.92458800E+03 Magnitude (dB) Calculated 20.6263744 -23.97775119 Magnitude (dB) SPICE 20.624635 -24.084018 Phase (deg) Calculated -89.04706182 -165.49354967 Phase (deg) SPICE -89.067371 -163.475026 Note: 1) SPICE analysis accounts for loop gain effects and closed loop phase shifts due to op amp Aol. 2) Calculated results do not account for loop gain effects and closed loop phase shifts due to op amp Aol. 33 Closed Loop Gain: Magnitude and Phase SPICE Ideal Op Amp & Poles: Equivalent Circuit DC Gain 100 VG1 f (Hz) ---------9.22017300E+01 3.92458800E+03 Buffer 1 RF 99kOhm + R1 1kOhm C1 15.9uF + + - - Magnitude (dB) Calculated 20.6263744 -23.97775119 Magnitude (dB) SPICE - Ideal 20.626374 -23.977751 CF 1.6nF + + - - Phase (deg) Calculated -89.04706182 -165.49354967 VOUT Phase (deg) SPICE - Ideal -89.047062 -165.49355 Note: 1) SPICE - Ideal Circuit analysis matches Calculated results. 2) No loop gain effect or closed loop phase shifts due to op amp Aol. 34 Closed Loop Gain: Magnitude and Phase SPICE Ideal Op Amp & Poles: Equivalent Circuit T 40 20 [B] 0 Gain (dB) SPICE Poles-Magnitude and Phase Ideal Op Amp Circuit [A] -20 Gain : -40 VOUT A:(92.20173; 20.626374) -60 Phase : VOUT A:(92.20173; -89.047062) -80 Gain : -100 VOUT B:(3.924588k; -23.977751) -120 Phase : -140 VOUT B:(3.924588k; -165.49355) -160 0 Phase [deg] -45 -90 -135 -180 1 10 100 1k 10k Frequency (Hz) 100k 1M 10M 35 3) Dual Feedback Paths and 1/b 36 Dual Feedback and 1/β Concept FB#1 RF Analogy: Two people are talking in your ear. Which one do you hear? The one talking the loudest! CF FB#2 RI - VO RO VOA + VIN VOUT Riso CL Dual Feedback: Op amp has two feedback paths talking to it. It listens to the one that feeds back the largest voltage (β = VFB / VOUT). This implies the smallest 1/β! + - 100 Dual Feedback Networks: Analyze & Plot each FB#? 1/β 80 A (dB) Aol Smallest FB#? dominates 1/β 60 1/β = 1 / (β1 + β2) FB#1 b FB#2 b 40 1/β relative to VO Note: VO = Op Amp Aol Output before Ro for this Dual Feedback Example 1/Beta [1/(b1 + b2)] 20 0 1 10 100 1k 10k Frequency (Hz) 100k fcl 1M 10M 37 Dual Feedback and 1/β How will the two feedbacks combine? Large b Answer: Small 1/β The Largest β (Smallest 1/β) will dominate! Small b Large 1/β - 38 Dual Feedback and the BIG NOT WARNING: This can be hazardous to your circuit! 100 Aol 80 A (dB) FB#1 1/Beta 60 FB#2 40 1/Beta 20 0 1 10 100 1k 10k 100k fcl 1M 10M Frequency (Hz) Dual Feedback and the BIG NOT: 1/β Slope changes from +20db/decade to -20dB/decade Implies a “complex conjugate pole ” in the 1/β Plot with small damping ratio, ζ. Implies a “complex conjugate zero” in the Aolβ (Loop Gain Plot) with small damping ratio, ζ. +/-90° phase shift at frequency of complex zero/complex pole. Phase slope from +/-90°/decade slope to +/-180° in narrow band near frequency of complex zero/complex pole depending upon damping ratio, ζ. Complex zero/complex pole can cause severe gain peaking in closed loop response. 39 Complex Conjugate Pole Phase Example From: Dorf, Richard C. Modern Control Systems. Addison-Wesley Publishing Company. Reading, Massachusetts. Third Edition, 1981. 40 Dual Feedback and 1/β Example RF 100k CF 82n FB#1 FB#2 VEE 12 VO 5V - U2 REF02 U1 OPA177E Vin Gnd REF02 + Trim Riso 26.7 Vout 5V + Out CL 10u VREF 5V VCC 12 Dual Feedback: FB#1 through RF forces accurate Vout across CL FB#2 through CF dominates at high frequency for stability Riso provides isolation between FB#1 and FB#2 41 Zo External Model for Dual Feedback Analysis RF 100k CF 82n VFB 4.999V VEE 12 - U2 REF02 U1 OPA177E Vin Gnd REF02 + Trim + Z o External Model VOA 4.9991V VCV1 Ro 60 x1 + + Out - - VO 4.9991V Vout Riso 26.7 4.9991V CL 10u VREF 4.999V VCC 12 Zo External Model: VCV1 ideally isolates U1 so U1 only provides data sheet Aol Set Ro to match measured Ro Analyze with unloaded Ro (largest Ro) which creates worst instability Use 1/β on Aol stability analysis 1/β, taken from VOA will include the effects of Zo, Riso and CL 42 Dual Feedback, FB#1 And FB#2 RF 100k CF 82n VFB 4.999V FB#2 VEE 12 - U2 REF02 U1 OPA177E Vin Gnd REF02 + Trim + Z o External Model VOA 4.9991V VCV1 Ro 60 x1 + + Out FB#1 - - VO 4.9991V Vout Riso 26.7 4.9991V CL 10u VREF 4.999V VCC 12 FB#1 and FB#2 1/ β Analysis: There is only one net voltage fed back as β to the -input of the op amp β_net = β_FB#1 + β_FB#2 This implies that the largest β will dominate → smallest 1/ β will dominate Analyze FB#1 with CF = open since it will only dominate at high frequencies Analyze FB#2 with CL = short since it is at least 10x CF 43 Dual Feedback and 1/β Example T 160 140 Aol 120 100 FB#1 1/b Gain (dB) 80 60 1/b FB#2 FB#1 fcl 40 20 fza fpc 1/b 0 fzx -20 Vout/Vin -40 10m 100m 1 10 100 1k Frequency (Hz) 10k 100k 1M 10M 44 Dual Feedback and 1/β – Create the BIG NOT VFB 4.999V LT 1T Aol = VOA FB#1: 1/Beta1= VOA / VF B Loop Gain = VFB RF 100k + CT 1T VG1 CF 220p VEE 12 Vtrim 1.2298V - U2 REF02 U1 OPA177E Vin Gnd REF02 + Trim + Zo External Model VOA 4.9991V Ro 60 VCV1 x1 + + Out - - VO 4.9991V Riso 26.7 Vout 4.9991V CL 10u VREF 4.999V VCC 12 45 Dual Feedback and 1/β – Create the BIG NOT T 160 140 Aol 120 100 FB#1 1/b Gain (dB) 80 60 1/b FB#2 FB#1 40 fcl 20 fpc fza 0 1/b -20 -40 10m 100m 1 10 100 1k Frequency (Hz) 10k 100k 1M 10M 46 Dual Feedback and 1/β – Create the BIG NOT T 160 140 Aol 120 100 ??? STABLE ??? Gain (dB) 80 60 fcl BIG NOT 40 20 BIG NOT 1/ 0 -20 -40 10m 100m 1 10 100 1k Frequency (Hz) 10k 100k 1M BIG NOT 1/b: At fcl rate-of-closure rule-of-thumb says circuit is stable but is it? 10M 47 Dual Feedback and 1/β – Create the BIG NOT ??? STABLE ??? BIG NOT Loop Gain: Loop Gain phase shift >135 degrees (<45 degrees from 180 degree phase shift) for frequencies <fcl which violates the loop gain phase buffer rule-of-thumb. But is it stable? 48 Dual Feedback and 1/β – Create the BIG NOT VFB 4.999V DC=0V Trans ient=10mVpp f =100Hz, 10ns rise/f all CF 220p VEE 12 Vtrim 1.2298V - U2 REF02 Vin Vin Gnd REF02 Trim Out RF 100k + + + Zo External Model VOA 4.9991V VCV1 Ro 60 x1 + + U1 OPA177E - - VO 4.9991V Riso 26.7 Vout 4.9991V CL 10u VREF 4.999V VCC 12 49 Dual Feedback and 1/β – Create the BIG NOT T 5.38 VO 4.34 6.10 VOA 3.04 100.00m Vin -100.00m 5.17 STABLE Vout 4.76 0.00 2.50m 5.00m Time (s) 7.50m 10.00m BIG NOT Transient Stability Test: Excessive ringing and marginal stability are apparent. Real world implementation and use may cause even more severe oscillations. We do not want this in production! 50 4) Non-Loop Stability Problems Non-Loop Stability • Loop Frequencies • RB+ • PCB Traces • Supply Bypass • Ground Loops • Output Stage Oscillations Non-Loop Stability Oscillations NOT predicted by Loop Gain (Aolb) Analysis or SPICE Simulations 52 Non-Loop Stability: Loop Frequency Definitions fcl: Where Loop Gain (Aolβ) = 1 Aol 100 fGBW: Where Op Amp Aol Curve crosses 0dB (Unity Gain Bandwidth) A (dB) 80 fGBW fcl 60 b 40 VOUT/VIN 20 SSBW (Small Signal BandWidth) 0 1 10 100 1k 10k 100k 1M 10M Frequency (Hz) 53 Non-Loop Stability: ? Diagnostic Questions ? Frequency of oscillation (fosc)? When does the oscillation occur? Oscillates Unloaded? Oscillates with VIN=0? 54 Non-Loop Stability: RB+ Resistor fosc < fGBW oscillates unloaded? -- may or may not oscillates with VIN=0? -- may or may not RF 100k RI 100k RB+ is Ib current match resistor to reduce Vos errors due to Ib. Ib - + VIN - VOUT Ib + RB+ 49.9k RB+ can create high impedance node acting as antenna pickup for unwanted positive feedback. + VNOISE PROBLEM RF - 100k RI 100k RF - 100k SOLUTION RI 100k VOUT + VIN - + RB+ 49.9k CB+ 0.01F or 0.1F If you use RB+ bypass it in parallel with 0.1μF capacitor. + VIN - - VOUT + RB+ SOLUTION Many Op Amps have low Ib so error is small. Evaluate DC errors w/o RB+. 55 Non-Loop Stability: PCB Traces fosc < fGBW oscillates unloaded? -- may or may not oscillates with VIN=0? -- may or may not RI RF + + Rs VIN - VOUT GND DO NOT route high current, low impedance output traces near high impedance input traces. Unwanted positive feedback path. DO route high current output traces adjacent to each other (on top of each other in a multi-layer PCB) to form a twisted pair for EMI cancellation. 56 Non-Loop Stability: Supply Lines fosc < fGBW oscillates unloaded? -- no oscillates with VIN=0? -- may or may not Ls PROBLEM Gain Stage PROBLEM Power Stage +vs RL IL - Rs + CL -vs Load current, IL, flows through power supply resistance, Rs, due to PCB trace or wiring. Modulated supply voltages appear at Op Amp Power pins. Modulated signal couples into amplifier which relies on supply pins as AC Ground. Power supply lead inductance, Ls, interacts with a capacitive load, CL, to form an oscillatory LC, high Q, tank circuit. 57 Non-Loop Stability: Proper Supply Line Decouple CLF SOLUTION CLF: Low Frequency Bypass < 4 in <10 cm RHF 10μF / Amp Out (peak) Aluminum Electrolytic or Tantalum CHF +VS < 4 in (10cm) from Op Amp - CHF: High Frequency Bypass + 0.1μF Ceramic -VS < 4 in <10 cm Directly at Op Amp Power Supply Pins CHF RHF RHF: Provisional Series CHF Resistance 1Ω < RHF < 10Ω CLF Highly Inductive Supply Lines 58 Non-Loop Stability: Ground Loops RI fosc < fGBW oscillates unloaded? -- no oscillates with VIN=0? -- yes RI RF RF + - - +VS VOUT VIN + -VS - RGw RL IL RGx + RGy “Ground” Ground loops are created from load current flowing through parasitic resistances. If part of VOUT is fed back to Op Amp +input, positive feedback and oscillations can occur. SOLUTION VIN PROBLEM RGv “Star” Ground Point - +VS VOUT + -VS RL RG Parasitic resistances can be made to look like a common mode input by using a “Single-Point” or “Star” ground connection. 59 Non-Loop Stability: Output Stages fosc > fGBW oscillates unloaded? -- no oscillates with VIN=0? -- no RF 100k L O A D RI 100k + VIN VOUT + RSN 10 to 100 - CSN .F to 1F -VS PROBLEM Some Op Amps use composite output stages, usually on the negative output, that contain local feedback paths. Under reactive loads these output stages can oscillate. SOLUTION The Output R-C Snubber Network lowers the high frequency gain of the output stage preventing unwanted oscillations under reactive loads. 60